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500W, CLASS E 27.12 MHz AMPLIFIER USING A SINGLE PLASTIC MOSFET
A P P L I C A T I O N N O T E
500W, Class E 27.12 MHz Amplifier Using A Single Plastic MOSFET
Richard Frey, P.E. Advanced Power Technology, Inc.
Bend, Oregon 97702 USA
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ABSTRACT
In this paper, we report on the design and evolution of a 500W, 27 MHz Class E amplifier. It doubles the operating frequency of previous high efficiency amplifiers using MOSFET transistors in the TO-247 package. Device criteria, circuit design, and amplifier performance characteristics are prented and compared to a HEPA computer model. INTRODUCTION
As the miconductor industry moves to larger size wafers to rai its productivity, they need more control of the processing. This translates into the need to produce a “harder” and more uniform plasma in vapor deposition and etching operations that rely on RF plasma. Typical operation is at 13.56 MHz but 27.12 MHz produces better results and is gaining incread popularity.
One of the challenges facing designers is finding devices suitable for rvice at the higher frequency. Around 1990, plasma equipment designers discovered that some of the inexpensive plastic high voltage MOSFETs they ud in their switchmode power supplies were capable of operating in high efficiency Class E rvice at 13.56 MHz. Since the switchmode devices operate at high voltage and cost much less than the alternative purpo-built RF parts, acceptance was almost immediate. Purpo-built RF devices operate on supply voltages below 50 Vdc. Suitable switchmode devices w
ith drain breakdown voltage of 1 kV can run on supply voltages up to 300 Vdc. They also demonstrated system level benefits operating at higher voltage. Power Combining and matching are easier since the drain impedance is higher. Lower RF current is less stress on the ries elements and enables the power supply design to be more efficient, smaller and lighter.
This RF application challenges most standard switchmode devices due to their packaging and die layout. The drain of the MOSFET is connected to metal back heat spreader, ud for heat sinking the device. This requires either an insulator between the package heat spreader and the heat sink or grounding the drain and changing the circuit topology to accommodate it. Either method adds asmbly cost of the PA and increas the potential for failures due to incorrect installation not to mention the poor thermal transfer characteristics of the insulator.
The package construction results in high source inductance, limiting the frequency respon of the devices. In some cas, the circuitous gate metallization layout caud gate signal propagation delay, limiting the frequency respon and reducing the effective power dissipation of the device. Raising the bar to 27 MHz reduces the number of suitable switchmode parts. This is primarily due to resistive loss associated with the polysilicon gate structure in the clod cell geometry ud in most switchmode devices. Input capacitance is esntially fixed by the device's power rating. Doubl
ing the operating frequency quadruples the gate dissipation. If the ESR of the gate capacitor is large, it cannot carry the RF current required to drive the device.
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The u of metal gate conductors dramatically reduces gate ESR. It is not unusual to reduce the gate resistance by a factor of 100 -- from 4 to 0.04ohms -- on devices of equal power rating. This complicates RF matching to the gate, but it is now rugged enough to operate reliably.
At 13 MHz, the device C OSS is combined with an external capacitor to obtain the required shunt output capacitance needed for proper class E operation [1].At 27 MHz, the C OSS is typically larger than that required for optimum Class E rvice. This compromis optimum Class E efficiency and output power capability. However, this would be a design consideration for any RF device, not just switchmode MOSFETs.
Figure 1: 27 MHz Class E Amplifier Schematic High voltage MOSFETs are now available that combi
ne the best practices from the RF world with the economy of the switchmode devices and packaging. They are available in mirror image pairs and the heat spreader of the plastic TO-247 package is connected to the source. They operate up to 300V V dd , at frequencies up to 100 MHz.AMPLIFIER DESIGN
The device ud for this amplifier is an APT ARF448A. It has a BV of 450V and R θJC of 0.55°C/W. The class C gain is more than 25dB at 27 MHz.C OSS is 125 pF at V dd =125V . R D(ON) is 0.4Ω. The data sheet parameters were entered in the HEPA [1]
*
PSPICE MODEL RF N-CHANNEL POWER MOSFET
*ARF448A/B 27 July 1998** G D S *.SUBCKT ARF448 6 4 1CISS 3 5 1450P CRSS 5 2 65P LG 7 6 4 6N
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M 8 5 3 3 125-050M L=2U W=1.4; DGSB LEVEL 1J1 8 3 2 125-050J D 3 2 125-050D LS 1 3 2 3N REGATE 7 5 .29LD 4 2 4 5N
.MODEL 125-050M NMOS (VTO=3.4 KP=14u
Lambda=1m Gamma=.2 RD=130m RS=13m)
.MODEL 125-050J NJF (VTO=-25.5 BETA=.01
女性体检项目一览表Lambda=.5)
.MODEL 125-050D D (BV=550 RS=230M CJO=422P
VJ=670M M=330M)
.ENDS
Figure 2. SPICE model for ARF448 MOSFET
C1,C375-380 pF mica trimmer, ARCO 465C4-C8.01 uF 1 kV disc ceramic C9,C10.1 uF 500V disc ceramic
L1 6 uH. 25t #am. 0.5" dia.L2210 nH. 4t #8 ga. .75" id, 1" long
金朝皇帝L42t #20 PTFE on .5" ferrite bead µ=850Q1APT ARF448A
R125Ω 5W non-inductive
T1
Pri: 4t #20 PTFE, Sec: 1t brass tube on 2 hole balun bead
Fair-Rite #2843010302 µ=850
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Table 1: 27 MHz Class E Amplifier Part Values
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Class E design program and it generated starting design values for the circuit. The circuit is common to most Class E amplifiers. [2] See Figure 1. In Class E amplifiers there is a capacitor shunted across the drain to source. There is none ud here becau the output capacitance of the device is slightly larger than the optimum value for that component. This defines the upper frequency limit for efficient Class E operation of a particular device. The 3Ω ESR of C OSS is one of the primary loss mechanisms in the circuit. However, C OSS exhibits little of the parasitic inductance that caud VHF ringing in the circuit described by Davis. The output circuit values were adjusted slightly for maximum efficiency at an output power of 490 W but basically, it worked "as adv
ertid" right off the bat.改过自新
While this description of the output circuit sounds straightforward, it was not reduced to practice very easily. The problem was providing enough rf voltage to the gate to drive the drain into saturation. The input impedance of the gate at 27 MHz is 0.1 -j2.7.C iss is 1400 pF. If 10V of peak gate drive is needed,a reasonable match between the drive source and gate is required. There is approximately 9 nH of parasitic gate inductance. This is enough inductance to make it impossible to obrve the actual voltage applied to "the gate" with an oscilloscope.SPICE was ud to model the gate drive circuit. A SPICE macro model for the device is shown in Figure 2. The goal was to design a network to match the gate impedance sufficiently to permit sine wave drive as in reference [2]. A circuit using a 4:1transformer and an L-network was designed using winSmith. [3] This worked, but the circuit was not stable. The parallel equivalent of the gate is 2200pF in parallel with 210 ohms. A 25 ohm 5W padding resistance was placed across the gate. This rais the effective input impedance to 0.38 -j2.6, lowers the network Q, and makes it much easier to match and drive properly.
The input transformer has been ud before [4]. It is made from a two hole ferrite "binocular" bead balun. The condary winding consists of two 7/8"pieces of 3/16" diameter brass tubing connected with
copper shim stock.
Figure 4: Drain voltage waveform of 500W
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Class E amplifier The four turn primary is wound inside the tubes for maximum coupling and minimum leakage which measured 19 nH referenced to the condary side.High quality passive components are required in the output network. Most important of the is L2. It was wound from #8 ga. bare copper wire, its Q measured 375, and the calculated dissipation is 4.2W. This coil is not capable of continuous duty operation unless it is attached with high temperature solder and/or parate mechanical termination support is ud. It was necessary to parallel three 10 nF Z5U ceramic coupling capacitors to carry the rf current.
COMPUTER SIMULATION
A simple SPICE model of the amplifier, similar to that ud by Davis in [2], was compared with HEPA results and the measured results of the operating amplifier. Overall, the agreement was good,especially between HEPA and the ideal circuit SPICE model. Attempts to inrt the SPICE macro model of the transistor into the amplifier was not successful. A much more sophisticated model is
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needed to adequately simulate the effects of the nonlinear capacitances of the MOSFET. However,the SPICE model was very uful for understanding the gate drive problem mentioned earlier.
Figure 3. Drain voltage waveform from HEPA HEPA assumes that the device is being driven into saturation, the only input parameter it considers is input drive power -- ud for the overall efficiency calculation. The gate drive was the biggest problem in the design becau a large RF-capable powe
r MOSFET has a very small input impedance.One of the best tests for reliability of an amplifier is mismatch load testing. Called "load pull" in some circles, it is a test which describes what the amplifier does when operating into a load other than 50 ohms.The ARF448 has about 175 watts of available dissipation in the test amplifier with its air cooled heat sink. The performance at eight points around a 2:1 VSWR mismatch circle was calculated using HEPA. Then the same test was run on the amplifier itlf with the drive duty cycle reduced to 50%. The various loads were obtained using a variable L-network and adjusting each load impedance with a vector impedance meter. The differences between the calculated and measured results were very small.The measured load pull results are shown in Figure 5. The most obvious conclusion is that there is a region in the high impedance inductive quadrant that should be avoided. Protection would be required to keep the transistor within its safe operating area if operated in this region.
Without protection, the output power soared to 750W and the drain current was almost twice normal.The efficiency stayed quite constant, never losing more than 11% at any load angle. Some form of
current foldback would be needed in the power supply of a practicable amplifier.
at 2:1 VSWR. Efficiency is displayed as % x 10.Class E amplifiers are ud commercially in RF sources of plasma generators and as exciters for CO 2lars. In plasma generator applications it is known that the load varies over a particular range. The feed line length is chon so the load prented to the amplifier does not enter the +45° quadrant. Other plasma generators have some form of active matching network between the amplifier and the load to handle the load pull issue. The drive or the drain voltage is reduced until a reasonable match is achieved. In CO 2 lars, the drive is pul width modulated and under high VSWR conditions, the pul width is reduced until the
load stabilizes.EXPERIMENTAL RESULTS
The efficiency of the Class E amplifier is 13% better with 30% more output power than was obtained from an initial Class C amplifier designed by classical methods using the same components. [5]The schematic did not change, only the values of L2 and C3 are different.
The performance of the amplifier is summarized in the table below along with the calculated values from HEPA. The results are fairly clo.
