With their low-input currents, FET input op amps are uni-versally ud in monitoring photodetectors, th
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e most com-mon of which are photodiodes. There are a variety of amplifier connections for this purpo and the choice is bad on linearity, offt, noi and bandwidth consider-ations. The same factors influence the lection of the amplifier with newer devices offering very low-input cur-rents, low noi and high speed.
西兰花的家常做法Photodetectors are the bridge between a basic physical indicator and electronics resulting in the largest single usage of FET op amps. As a measure of physical conditions, light is condary to temperature and pressure until the measure-ment is made remotely with no direct contact to the moni-tored object. Then, the signals of a CAT scanner, star-tracking instrument or electron microscope depend on light for the final link to signal processing. Photodiodes have made that link economical and expanded usage to detector arrays that employ more than 1000 light nsors. Focus then turns to accurate conversion of the photodiode output to a linearly related electrical signal. As always, this is a contest between speed and resolution with noi as a basic limiting element. Central to the contest is the emingly simple current-to-voltage converter which displays surprising mul-tidimensional constraints and suggests alternative configura-tions for many optimizations.
CURRENT-TO-VOLTAGE
The energy transmitted by light to a photodiode can be measured as either a voltage or current output. For a voltage respon, the diode must be monitored from a high imped-ance that does not draw significant signal current. That condition is provided by Figure 1a. Here, the photodiode is in ries with the input of an op amp where ideally zero current flows. That op amp has feedback t by R
1 and R
2 to establish amplification of the voltage diode just as if it was an offt voltage of the amplifier. While appealing to more common op amp thinking, this voltage mode is nonlinear.The respon has a logarithmic relationship to the light energy received since the nsitivity of the diode varies with its voltage.
Constant voltage for a fixed nsitivity suggests current output instead and that respon is linearly related to the incident light energy. A monitor of that current must have zero input impedance to resp
ond with no voltage across the diode. Zero impedance is the role of an op amp virtual ground as high-amplifier loop gain removes voltage swing
from the input. That is the key to the basic current-to-voltage converter connection of Figure 1b. It provides an input resistance of R 1/A where A is the open-loop gain of the op amp. Even though R 1 is generally very large, the resulting input resistance remains negligible in comparison to the output resistance of photodiodes.
FIGURE 1a. Photodiode Output Can be Monitored as a
Voltage; or, 1b, as a Current.Diode current is not accepted by the input of the op amp as its prence stimulates the high amplifier gain to receive that current through the feedback resistor, R 1. To do so, the amplifier develops an output voltage equal to the diode current times the feedback resistance, R 1. For that current-to-voltage gain to be high, R 1 is made as large as other constraints will permit. At higher resistance levels, that resistor begins to develop significant thermal DC voltage PHOTODIODE MONITORING WITH OP AMPS托马斯英语
a respon zero due to C D and begins a ri that is terminated only becau of a cond parasitic capacitance. Stray capaci-tance, C S , shunts the feedback resistor resulting in a respon pole leve
ling the gain at 1+ C D /C S . For large area diodes C D can be hundreds of picofarads causing the noi gain to peak in the hundreds as well. That gain continues to higher frequencies until rolled off by the op amp bandwidth limit.As feedback resistance increas, the pole and zero of this gain peaking move together to lower frequencies encom-passing a greater spectrum with high gain.
First signs of this gain peaking phenomena are familiar to anyone who has ud high resistance op amp feedback in more general circuits. High output to input resistance with an op amp results in overshoot, respon peaking, poor ttling or even oscillation all due to the resistance interac-tion with amplifier input capacitance. Together the resis-tance and capacitance form another pole in the feedback loop resulting in the classic differentiator feedback respon.Shown by the dashed line for more general op amp cas, the associated feedback factor reciprocal intercepts the ampli-fier open loop magnitude respon with a 12dB/octave rate
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FIGURE 2a.Due to Diode Capacitance in the Feedback of
the Basic Current-to-Voltage Converter, 2b,Op Amp Noi Receives Gain and Bandwidth Not Available to the Signal.
drift due to the temperature coefficient of the amplifier input current. To compensate this error, an equal resistance R 2 is commonly connected in ries with the op amp noninverting input, as shown, and capacitively bypasd to remove most of its noi. The remaining DC error is determined by the mismatches between the amplifier input currents and be-tween the two resistors. A drawback of this error correction is the voltage drop it creates across the diode and the resulting diode leakage curren
t. That leakage can override the correction achieved with R 2, as photodiodes typically have large junction areas for high nsitivity. Leakage cur-rent is proportional to that area which can become much larger than the op amp input currents.
Only zero diode voltage can eliminate this new error source but that is in conflict with control of a cond attribute of large diode area. Large parasitic capacitance is also prent creating often vere amplification of noi as will be described. To reduce that capacitance, a large rever-bias voltage is sometimes impresd on the diode greatly compli-cating DC stability and making current noi from the photodiode an additional error factor. Larger diode area may actually degrade overall accuracy and higher photo nsitiv-ity should first be sought through optical means such as a package with an integral molded lens. Monitor-circuit con-figurations that maintain zero diode voltage are also candi-dates in this optimization and are described with Figures 6,7 and 9.
The value of the feedback resistor in a current-to-voltage converter largely determines noi and bandwidth as well as gain. Noi contributed directly by the resistor has a spectral density of √4KTR 1 and appears directly at the output of a current-to-voltage converter without amplification. Increas-ing the size of the resistor not only rais output noi by a square root relationship but also increas output signal by a direct proportionality. Signal-to-noi ratio, then, tends to increa by th
e square root of the resistance.
Noi from the op amp also influences the output with a surprising effect introduced by high feedback resistance and the diode capacitance. The amplifier noi sources are mod-eled in Figure 2a as an input noi current, i n, and the input noi voltage, e n . The current noi flows through the feed-back resistor experiencing the same gain as the signal current. It is the shot noi of the input bias current, I B , and has a noi density of √2qI B 1. Choice of an op amp having input currents in the picoamp range makes this noi com-ponent negligible for practical levels of feedback resistance.Input noi voltage of the amplifier would at first em to be transferred with low gain to the output. That is true at DC where its gain 1 + R 1/R D is kept small by the large diode resistance, R D . Capacitance, C D , of the diode alters the feedback at higher frequencies adding very significant gain to e n . As both the capacitance and the feedback resistance are commonly large, the effect can begin at fairly low frequen-cies. Figure 2b illustrates the effect with an op amp gain magnitude curve plotted with the reciprocal of the feedback factor or the “noi gain.” That gain curve first experiences
wide bandwidth first encompass the peaking. The noi curves level off when esntially the full amplifier band-width is encompasd by the gain peaking. Moving to yet higher resistance, resistor n
oi would return the curves to rising slopes, but resistor bandwidth is by then rolled off by stray capacitance. In this upper region, any increa in resistance is accompanied by a matching reduction in noi bandwidth so that the total resistor noi becomes a constant.Variables of diode and stray capacitances alter the point of ont of gain peaking errors, but the characteristic shape of the output noi curves remains the same for any ca. Each will display ranges dominated by op amp noi, resistor noi and gain peaking effects.
FIGURE 3. As the Feedback Resistance of a Current-to-Voltage Converter Increas, the Dominant Noi Source Changes from the Op Amp to the Resistor and Back to the Op Amp under Gain Peaking Conditions.Comparing the curves shows that the OPA111/OPA2111provide the lowest noi in two of the characteristic ranges.While the OPA128 shows a lower noi curve in the middle range, that is due to the amplifier’s lower bandwidth and a bandwidth reduction technique to be described, removes that difference for the OPA111. Where the OPA128 excels is in very low DC error as its input currents are a mere 0.075pA which is 1/20th that of its low-noi contender. The third op amp, OPA404, produces higher total output noi overall,but that again is largely a bandwidth phenomenon. The 6.4MHz respon of that amplifier accommodates noi over a much greater frequency range. While the noi curve for this amplifier is consistently higher than that of the OPA1
28, the OPA404 actually has lower noi density but it has six times the bandwidth. That 6.4MHz bandwidth is available to signals for feedback resistances up to 50k Ω and the amplifier still offers the best bandwidth for resistances up to 150k Ω. As the OPA404 is a quad op amp, its economy suggests consideration for u at even higher resistances along with bandwidth reduction that provides more competi-tive output noi.
of closure corresponding to feedback pha shift approach-ing or equal to 180°. The common cure for this condition is a capacitor across the feedback resistor, which for the very high resistances of cu
rrent-to-voltage converters, automati-cally results from stray capacitance. Such capacitance de-generates the added feedback pole to control pha shift in the feedback loop.
In understanding current-to-voltage converter noi perfor-mance it is important to note that the signal current and the noi voltage encounter entirely different frequency re-spons. The current-to-voltage gain is flat with frequency until the feedback impedance is rolled off by stray capaci-tance as shown. Gain received by the amplifier noi volt-age, on the same graph, extends well beyond that roll-off and is high in that extended region. The majority of the op amp’s bandwidth often rves only to amplify that noi error and not the signal. This is typically the dominant source of noi for higher feedback resistances.
Relative effects of the major noi sources of a current-to-voltage converter can be en with the curves of Figure 3.Tho curves show output noi for the basic current-to-voltage converter of Figure 1b including the effects of the noi gain reprented in Figure 2b. Plotted are total output nois for three cas as a function of feedback resistance and each is the rms sum of the components produced by the feedback resistor and an op amp. Reprented are three FET op amps having different performance specialties that cover the spectrum of photodiode applications with low noi,low-input bias current and high speed. While all three types have low-noi designs a
nd low-input currents, the OPA111offers the lowest noi in the FET op amp class at 6nV/√Hz,and the OPA128 has the lowest input current at 0.075pA.Without neglecting performance in the categories, the OPA404 design pushes bandwidth to 6.4MHz. Noi due to the op amp is found by integrating the amplifier noi density spectral respon over the noi gain respon 2. Also shown, by a dashed line, is the noi due to the resistor alone for the OPA111 and OPA2111 ca. This resistor noi curve is different for the other op amps as each amplifier has a different bandwidth rolling off noi due to the resistor.Different factors control the noi curves for different ranges of feedback resistance. At low resistance levels, the noi curves are largely flat with the op amp voltage noi the dominant contributor. That domination makes initial resis-tance increas have little effect except for the ca of the very low-voltage noi of the OPA111/ OPA2111. In this region noi gain peaking has not yet been encountered so the output noi remains small. Between 10k Ω and 1M Ω,resistor noi is dominant and the curves track that error source as the dashed line shows for the OPA111/OPA2111.Here, the curves demonstrate the square root relationship with the resistance and differ only becau of amplifier bandwidths. At still higher resistance, noi gain peaking takes effect returning the op amp noi to dominance and boosting the curves higher. That effect is first demonstrated by the incread slope of the OPA404 curve as that amplifier’s
Only a five dimensional graph could display the output noi, resistance, DC error, diode area and signal bandwidth considered in current-to-voltage converter design. Each spe-cific application’s requirements are evaluated parately with respect to the factors. To avoid suboptimizing a given design for one factor such as gain, the various effects of increasing feedback resistance are anticipated at each step.Choices such as large diode area are made considering the related capacitance and its effect on output noi and overall circuit nsitivity.
NOISE CONTROL
Gain peaking effects are the primary noi limitation with the commonly preferred high feedback resistances. To limit this effect, or to eliminate the gain ri entirely, additional capacitance is commonly added to bypass the feedback resistor. The capacitance level required can be very small for some values of R 1 and the relative significance of unpredict-able stray capacitance make tuning desirable. Combined,the requirements are a challenge better resolved with a capacitor tee network as described in Figure 4a. It is capable of even subpicofarad tunable capacitance with little effect on stray capacitance in the tuning operation. The tee us a capacitive divider formed with C 2 and C 3 to attenuate the signal applied to C 1 at the circuit input. With only a fraction of the output signal on C 1, it supplies far less shunting current to the input node as would a much smaller capacit考试祝福
or.Controlling the attenuation ratio is the tunable C 3, which is the largest of the capacitors, so its capacitance value is more readily available in tunable form. Since that capacitor is grounded, it has a shielding advantage to reduce stray capacitance influence while tuning.
出游方案Another option for practical feedback bypass exists with a resistor tee which is a commonly considered replacement for the high value feedback resistor. The latter is replaced in Figure 4b by elements of more reasonable value but intro-duces greater low frequency noi. Its operation is the dual of the capacitor tee above with R 2 and R 3 attenuating the signal to R 1 so that the latter appears as a much larger resistor to the input node. A similar opportunity for the DC error compensation resistor R 2 does not exist. DC error due to amplifier input current is no different with the tee so the large compensation resistor is still needed.
Stray capacitance across the feedback is somewhat reduced with the resistor tee by the added physical spacing of the feedback with three elements. Also, stray capacitance across each individual element has much less effect with their lower resistances. Sensitivity to other stray capacitance from the op amp output to its input has the same effect as before.In the attenuation network of the feedback is the opportunity for intentional bypass with reasonable capacitor values.Bypassing the moderate resistance of R 2 removes the attenu-ation at higher frequencies leaving the net feedback
resis-tance at the level of R 1. This operation differs from true feedback bypass in that impedance levels off, rather than continuing to fall with frequency, but the dramatic drop in
FIGURE 4a.Removal of Amplifier Gain Peaking Through
Small Capacitive Bypass of Large Feedback Resistance is More Feasible with a Capacitor Tee; or, 4b, Bypass of One Element of a Feed-back Resistor Tee.equivalent resistance rves the circuit requirement. Another benefit offered by the resistor tee is more accurate DC error compensation.
Reduced high frequency noi with the tee element bypass is accompanied by an opposing increa at lower frequen-cies. Below the frequency of the bypass, noi gain is incread by the feedback attenuation of the tee network.That amplifies the noi and offt voltages of the op amp as well as the noi of resistor R 1 by a factor of 1 + R 2/R 3.Countering the latter is the resistor’s smaller value so that
shown to continue its attenuating amplifier action well beyond the unity gain crossover of A 1. This avoids a cond gain peak that could cau oscillation. Signal bandwidth of the current-to-voltage conversion is esntially unaffected as R 1 has not been influenced .
Where the Figure 5 technique is most uful is with lower level signals that have greater nsitivity to noi. In higher level applications that circuit can encounter a voltage swing limitation but another u of the cond amplifier offers similar noi improvement. The swing limitation results from the maximum output voltage limit of A 1 and its attenuation by A 2. If the output of A 1 has a peak swing of 12V and A 2 has the gain of –1/10 illustrated, the final output is limited to a 1.2V peak swing. For lower-level signals this will be acceptable as the maximum practical level of feed-back resistance already limits output swing.
Higher-level signals are not as nsitive to noi and better tolerate a more straight forward approach to filtering. An active filter following the conventional current-to-voltage converter also removes the high frequency noi. Setting filter poles at the frequency of the signal bandwidth results
this effect is incread only by the square root of the new noi gain. Most important, however, is the bypass capacitor removal of high frequency gain as it eliminates the greatest portion of previous noi bandwidth. In the abnce of other means to remove the high frequencies, the bypasd resistor tee provides lower total output noi for the higher ranges of feedback resistance.
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Adding feedback capacitance is an effective means of reduc-ing noi gain but it also decreas signal bandwidth by the same factor. That bandwidth is already low with high feed-back resistance and the end result can be a respon of a kilohertz or less. A more desirable solution to the noi problem is to limit amplifier bandwidth to a point just above the unavoidable signal bandwidth limit. Then, the high frequency gain which only amplifies noi is removed. Op amps with provision for external pha compensation offer this option, but tho available lack the low-input currents and low-voltage noi needed for photodiode monitoring.To achieve this bandwidth limiting with better suited op amps, a composite amplifier us two op amps with the added one for pha compensation control as in Figure 5a.Note the reversal of the inverting and noninverting inputs of A 1 needed to retain a single pha inversion with two amplifiers in ries. With the composite structure, internal feedback controls the frequency respon of the gain added by A 2. At DC, that feedback is blocked by C 1 and overall open-loop gain is the product of tho of the two amplifiers or 225dB for tho shown. That gain is rolled off by the open-loop pole of A 1 and by the integrator respon estab-lished for A 2 by C 1 and R 3. As this is a two pole roll-off, it must be reduced before intercepting the noi gain curve to establish frequency stability. A respon zero does this due to the inclusion of R 4. Above the frequency of that zero, R 4also replaces the integrator respon with that of an inverting amplifier having a gain of –R 4/R 3. Making that gain less than unity drops the net
gain magnitude curve below that of a single amplifier at high frequencies. Graphically, the noi gain respon of Figure 5b is moved back in frequency much as if the op amp bandwidth had been reduced.
Eliminated is the shaded area of noi gain, which visually may not appear dramatic, but that is becau of the logarith-mic-frequency scale. Actually, the associated noi reduc-tion is large becau most of the amplifier’s bandwidth is reprented in this upper end of the logarithmic-respon curve. Moving the unity gain crossover of the noi gain from 2MHz to 200kHz, as shown, drops the output noi due to A 1 by about a factor of three. To achieve the same result with feedback bypass, the signal bandwidth would have been reduced a factor of ten. That bandwidth is unaffected with the Figure 5a approach. No noi, or offt, is added by A 2 as this amplifier is preceded by the high gain of A 1. With the exceptionally low noi of the OPA111 input amplifier,this improvement reduces noi to the fundamental limita-tion impod by that of the feedback resistor. This condition is retained for all practical levels of high feedback resis-tance. For the cond amplifier, the wideband OPA404 is
FIGURE 5a. Noi Reduction Results with a Composite
Amplifier that, 5b, Restricts Noi Bandwidth Without Reducing that of the Signal.
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