A NEW APPROACH TO BROADBAND ARRAY DESIGN
USING TIGHTLY COUPLED ELEMENTS
Mark Jones and James Rawnick
Harris Corporation
Melbourne, FL
ABSTRACT
A fundamentally different approach to broadband array design has been developed and validated with measured hardware. Traditional array design is bad on designing an isolated element with the desired bandwidth and radiation characteristics and does not account for the mutual coupling that occurs when this element is placed in an array environment. It is well known that undesired mutual coupling in an array can cau dramatic changes in antenna element impedance and radiation patterns. Harris’ new approach treats the array aperture as a periodic surface so that the performance of elements within the array environment is assured. The approach is similar to that ud to design a frequency lective surface, another periodic, highly-coupled structure. The resultant array apertures a
re inherently low-profile in cross-ction, amenable to conformal implementations, and are able to maintain efficient characteristics over large scan volumes with bandwidths approaching a decade. This paper will prent an introduction to this design methodology supported by predicted and measured results from veral array designs.
INTRODUCTION
Phad array design has been historically complicated by the prence of mutual coupling between elements which have been optimized in isolation. Mutual coupling effects are typically manifested in performance metrics such as impedance bandwidth or axial ratio bandwidth. Various approaches have been utilized to suppress undesired coupling between array elements such as cavities or cups behind each element, conductive traces around each element, and surface wave absorber between elements1,2,3. The and other methods have met with varying levels of 1Cox, Graham G., “Development of L-band Feed Arrays for Inmarsat-3,” IEE Colloquium on Inmarsat-3, September 24, 1991.
2Rawnick, et al., Suppression of Mutual Coupling in an Array of Planar Antenna Elements. US Patent 6,583,766, June 24, 2003.
3Wren, Lloyd. Microwave Absorber. US Patent 4,381,510, April 26, 1983.
1-4244-1513-06/07/$25.00 ©2007 IEEE success and may involve a reduction in gain or an increa in complexity.
The design of planar phad arrays having large scan volumes and bandwidths approaching a decade prents an even greater challenge due to the small element spacing that is required to prevent the introduction of grating lobes. An array lattice of one-half wavelength allows the main beam to be scanned over all space without grating lobes4. The appearance of a grating lobe can be accompanied by a reduction in gain or an impedance discontinuity for arrays with significant inter-element coupling. For the reasons, it is generally recommended that a spacing which permits grating lobes be avoided in array design5.
Elements which exhibit desirable characteristics over a 10:1 frequency band and therefore may be lected to form broadband arrays are usually spirals, conical helices, tapered slots, or log-periodic elements. The elements may be too large for the required array lattice or may not be suitable for planar implementations. The classical design process of designing a broadband element and attempting to mitigate the effects of mutual coupling inherent to the array environment has prevented
the development of broadband phad arrays which are capable of large scan volumes and suitable for conformal applications. The inability to analyze the effects of mutual coupling becau of the required computational resources has also contributed to the continued usage of the classical technique. Meanwhile, the demand for the types of systems in the military arena is steadily increasing due to the desire to replace multiple nsors with a single multi-function aperture in a low-profile installation, provide higher data rates, and operate over a wide field of view.
THE CURRENT SHEET ARRAY CONCEPT Harris Corporation has developed a new approach to broadband array design in order to remove the limitations encountered in the conventional design process. This has resulted in a class of broadband, low-profile apertures
4Mailloux, Robert J. Phad Array Antenna Handbook, Artech Hou, 2005, pp. 27-30.
5Milligan, Thomas A. Modern Antenna Design, Wiley-IEEE Press, 2005, pp. 125-133.
which exhibit well-behaved impedance and radiation patterns over bandwidths approaching a decade, can provide single or dual polarization, are well-suited for conformal implementations, and exhibit high cross-polarization isolation. The approach consists of applying broadband impedance matching techniques to an array of tightly-coupled, electrically small elements. The inter-element cou
pling is introduced on the aperture layer and should be distinguished from the mutual coupling typically measured at the element outputs. The idea of using capacitively-coupled dipoles embedded in stratified dielectric layers above a ground plane was conceived by Dr. Ben Munk of The Ohio State University and was reduced to practice at Harris6. Extending the tightly-coupled element approach to orthogonal dipoles yields a dual-polarized aperture that rves as a practical implementation of Wheeler’s theoretical “current sheet” concept7 with a ground plane below the radiating surface. The methodology of forming broadband arrays from tightly-coupled, electrically small elements has been proven by other rearchers to be applicable to spirals8, pixelated elements9, and tapered slots10. Harris has demonstrated that the Current Sheet Array (CSA) technology can be applied to a wide range of frequency bands and array sizes. The results of this groundbreaking development reprent the current state of the art in broadband array design.
THEORY OF OPERATION
It has been shown that arrays of dipole elements, even tho in clo proximity to a ground plane, can maintain good radiating efficiency provided that the input impedance can be matched11. Historically, this has proven difficult to implement over a wide range of frequencies due to the electrically small ground plane spacing at the low end of the band. The CSA approach address thi
s 6Taylor, et al., Wideband Phad Array Antenna and Associated Methods. US Patent 6,512,487, January 28, 2003.
7H.A. Wheeler, “Simple relations derived from a phad-array antenna made of an infinite current sheet," IEEE Trans. Antennas Propagat., Vol AP-13, pp. 506-514, July 1965.
8Riddle, et al., Conformal, Low RCS, Wideband, Phad Array Antenna for Satellite Communications Applications. US Patent
6,300,918. October 9, 2001.
9Maloney, et al., Fragmented Aperture Antennas and Broadband Antenna Ground Planes. US Patent 6,323,809. November 27, 2001.
10Kragalott, et al., “Design of a 5:1 Bandwidth Stripline Notch Array from FDTD Analysis,” IEEE Trans. Antennas Propagat., Vol AP-48, No. 11, pp. 1733-1741, November 2000.
11J. L. Allen, “Gain and Impedance Variation in Scanned Dipole Arrays,” IRE Transactions on Antennas and Propagation, Sept. 1962 , pp. 566-572.fundamental broadband impedance matching problem in order to achieve a broadband radiating aperture.
The theoretical development of the CSA is bad upon analyzing the impedance behavior of each component of the design in order to construct the total impedance respon of the array. Appropriate lection of the design elements, including element type and coupling, ground plane spacing, and dielectric matching layers can result in nearly constant impedance over a wide frequency range even though the constituent impedance terms vary over frequency. An entire chapter of the text by Munk is dedicated to providing an explanation of this procedure12. Equivalent circuits are ud to demonstrate the effect of each constituent of the array and lead to an intuitive feel for the array behavior. Assumptions that are made in order to utilize this approach include that the elements are electrically short, the array is infinite, no grating lobes are allowed in real space, and scan performance is considered for principal planes only. Given the assumptions, the circuit model is an exact equivalence for the impedance respon of the array.
An overview of this analysis is prented herein and begins by considering the equivalent circuit of an array of coupled dipoles spaced above a ground plane with a single dielectric layer above the elements. This is a reprentative profile that can be ud to demonstrate the analytical behavior of the CSA, although various designs may differ from this example and are generally optimized using electromagnetic design tools. The geometry and equivalent circuit are shown in Figure 1, using nom
enclature consistent with the Munk text. The impedance at the antenna terminals is Z A = jX A1+ Z1- || Z1+, where the array in free space has the terminal impedance Z A0 = R A0+jX A0, and in an infinite dielectric medium has the terminal impedance Z A1 = R A1+jX A1. Z1+ is the impedance viewed from the array terminals towards the ground plane, and Z1-is the impedance viewed from the array terminals towards free space through the dielectric cover layer. X A1 is the terminal reactance of the dipole array. Note that the impedances viewed from the array terminals combine in parallel since the values are transformed along the appropriate electrical distances to be coincident at the array terminals.
12Munk, Ben A. Finite Antenna Arrays and FSS, Wiley-IEEE Press, 2003, pp. 181-213.
Figure 1(a). Example array geometry.
Figure 1(b). Example equivalent circuit.
Figure 2. Impedance of ground plane as viewed from array
terminals.
As shown in Figure 2, the impedance of the ground plane viewed from the array terminals Z 1+ varies with frequency. For a reference frequency f 0, the array is spaced an electrical distance of λ/4 away from the ground plane. It is well known from transmission line theory that the short
circuit zero impedance of a ground plane is transformed to
浇汁豆腐
an open circuit infinite impedance over an electrical
distance of one-quarter wavelength. The impedance of the
ground plane travers along the outer edge of the Smith
chart since it is purely reactive, and appears inductive for frequencies below f 0 and capacitive for frequencies above f 0. Therefore the lection of an element with the proper reactance variation with frequency can counteract the reactance variation of the ground plane. Since the
reactance of a dipole is capacitive for lower frequencies and inductive for higher frequencies, it is an ideal choice to reduce the overall impedance variation of the array and ground plane combination. This property alone can be exploited to yield a bandwidth of veral octaves from an array of dipoles spaced above a ground plane 13.
The u of dielectric layers can further increa bandwidth by providing additional cancellation of the ground plane variation with frequency. As shown in Figure 3, the impedance viewed through the single dielectric layer above the dipole array also varies with frequency, but remains on the circle passing through the center of the Smith chart and through the real impedance Z 1- = (2R A1)2/2R A0 at f 0, the frequency where the dielectric electrical thickness is λ/4. For this frequency, the dielectric rves as a quarter-wavelength impedance transformer with an impedance 2R A1 to transform the impedance 2R A0 to the value Z 1-. Note that the impedances are referenced to 2R A0, which is the value at the center of the Smith chart. The primary effect of the dielectric layer is to transform the impedance of free space through the dielectric to produce a smaller value for Z 1-, which yields a greater bandwidth when combined in parallel with the groundplane reactance Z 1+. The parallel combination of Z 1- and Z 1+ is found by adding the pure susceptance of Y 1+ to the complex admittance Y 1-. As illustrated in Figure 4, this can be performed on an admittance Smith Chart by ta
king each point on the Z 1- curve and moving along the circle passing through that point and the Smith Chart origin a distance equal to the value of Z 1+ for that
frequency.
Figure 3. Impedance of free space transformed through a dielectric layer as viewed from array termi
nals.
13 Munk, Ben A. Finite Antenna Arrays and FSS, Wiley-IEEE Press,
2003, pp. 186-187.
Figure 4. Impedance of parallel combination of Z1- and
Z1+ as viewed from array terminals.
藕的营养价值Figure 5. Final impedance at array terminals. Finally, the array terminal impedance Z A is found as shown in Figure 5 by adding the reactive component of the dipole array jX A to Z 1- || Z 1+. Since jX A is capacitive for frequencies below f 0 and inductive for frequencies above f 0, the outer points of the Z 1- || Z 1+ locus are drawn in toward the real axis. This Smith chart illustrates how the combination of the dipole array, ground plane, and dielectric layers can achieve a broadband impeda
nce match. The dielectric layers can also be employed to stabilize the impedance variation as the array beam is scanned from broadside.
As the array lattice decreas, the real part of the dipole array terminal impedance increas since it is approximately inverly proportional to the element spacing 14. For short element lengths it can approach veral hundred Ohms, and this has been verified for the CSA. This is in contrast to the isolated dipole resistance which decreas as the element length becomes smaller 15. In order to understand why capacitive coupling is utilized between the elements, consider that a dipole array at resonance has an equivalent inductance near the terminals due to the wires and an equivalent capacitance near the
14
Mailloux, Robert J. Phad Array Antenna Handbook, Artech Hou,
2005, p. 323.
15
Johnson, R.C. and Jasik, Henry. Antenna Engineering Handbook, McGraw-Hill Book Co., 1984, Cha
凶猛的意思pter 4.
ends due to fringing fields. As the element length decreas, the elements become less inductive and have less end capacitance, so the resonant frequency increas. Therefore more capacitance is added between the ends to maintain the condition of zero reactance.
HARDWARE IMPLEMENTATION吃的笔顺>正规借条模板
The initial design of the CSA was performed using the Periodic Moment Method (PMM) code, which can analyze infinitely periodic planar structures comprid of thin wires and lumped circuit elements surrounded by dielectric layers 16. This tool was originally developed for the design of passive periodic structures known as frequency lective surfaces (FSS) and was later expanded to include arrays of active elements. Commercial full-wave three-dimensional electromagnetic field solvers such as Ansoft HFSS and CST Microwave Studio have been ud to perform detailed design trades and have yielded good agreement with measurements.
The predicted broadside impedance from PMM for an example single-polarization design is shown in Figure 6 and demonstrates the broadband nature of the CSA concept. A short ries matching ction known as a “pigtail” was included in order to further compress the impedance curve. This ma
tching ction may be
implemented using various methods.
Figure 6. Predicted impedance from PMM for an example
CSA design with a short matching ction. The implementation of the inter-element coupling was accomplished using interdigital capacitors as shown in Figure 7. The array shown is an early single-polarization breadboard targeted for the 2-18 GHz band. The interdigital capacitor allows a wide range of coupling values to be obtained from a single layer printed wiring
16
L.W. Henderson, “Introduction to PMM," Tech. Rept. 715582-5,
OSU ESL, Dept. of Electrical Eng., prepared for Wright-Patterson Air Force Ba, OH, Feb. 1986.
board. Later investigations led to alternate embodiments for the coupling between the ends of the elements 17,18
.
Figure 7. Early single-polarization CSA breadboard showing interdigital coupling between elements.
The implementation of the element feed was realized using a machined device known as a “feed organizer” to route
coaxial cables to the elements 19. A typical feed organizer
for a dual-polarization array is shown in Figure 8. It was
determined that it was necessary to prevent common-mode
如何养虾
currents from flowing on the feed cables and corrupting the array performance. The feed organizer provides
common-mode suppression and a reliable connection from
the feed components to the CSA elements.
Figure 8. Dual-polarization feed organizer. Figure 9 shows a dual-polarization breadboard designed
for 2-18 GHz which contains an 8x8 connectorized ction
17
Durham, et al., Phad Array Antenna with Discrete Capacitive
Coupling and Associated Methods. US Patent 6,856,297, February 15, 2005. 18
Durham, et al., Multi-Layer Capacitive Coupling in Phad Array Antennas. US Patent 6,822,616, November 23, 2004. 19
Rawnick, et al., Patch Dipole Array Antenna Including a Feed Line Organizer Body and Related Methods. US Patent 6,483,464, November 19, 2002.
embedded in an array of 2,664 elements. Commercially available 0°/180° hybrids were ud to provide the correct phasing to the dipole arms. Elements that were not connectorized were terminated with resistive loads. The array size of 22” x 22” demonstrated that a large array could be manufactured as a single unit, however it is larger than necessary to terminate the active region. Typically the active region is surrounded by only a few rows of resistively terminated elements, which is similar to edge termination techniques ud for other types of arrays. The spacing from the elements to the ground plane is less than λ
/20 at the low end of the band.
Figure 9. Aperture and feed network views of dual-polarization CSA breadboard designed for 2-18 GHz.
Figure 10 shows the measured broadside gain for the 8x8
active region of this breadboard from 2-18 GHz. This is compared to the theoretical maximum directivity achievable from an equivalent area within an infinite planar array 20. Excellent agreement is obtained between the measurement and the HFSS prediction. A single unit cell was modeled in HFSS with periodic boundaries to reproduce the unit cell in an infinite array environment. The predicted unit cell gain was then incread by 10*log(64) to calculate the predicted gain for the 8x8 array. The decrea in gain at the high end of the band is caud by the electrical distance between the elements and the ground plane nearing λ/2. As expected, the array will
exhibit a radiation null at broadside for this ground plane spacing.
20
Balanis, Constantine. Antenna Theory: Analysis and Design, John Wiley & Sons, Inc., 1997, pp. 84-86.
台湾通行证办理流程Figure 10. Measured vs. predicted gain of 8x8 active
region of CSA breadboard in Figure 9. The impedance and VSWR for a centrally-located element in t
he 8x8 active region is shown in Figure 11. The VSWR is 3:1 or better over the entire 2-18 GHz band, and better than 2:1 over much of the band. The impedance is fairly constant with frequency and is referenced to the 50 Ω input port of the hybrid ud to feed each element. The actual feedpoint impedance of the element is twice the impedance shown in Figure 1121
.
Figure 11. Measured impedance of centrally-located
element from CSA breadboard in Figure 9. E-plane and H-plane radiation pattern cuts of the 8x8 active array are given in Figure 12 for 2 GHz and 18 GHz. The array patterns were well-behaved and showed high cross-polarization isolation. The measured cross-polarization coupling between orthogonal dipole elements was better than -30 dB across the band. The array was
21
Davis, W.A., Nealy, J.R., Ricciardi, G.F., and Stutzman, W.L.,
“Techniques for the measurement of the impedance of wideband balanced antennas,” IEEE AMTA Symposium, Nov. 1995.
scanned using true time delays and good scan performance was achieved.美国艺术留学
As with other coupled broadband arrays, careful design of the lattice, feed implementation, and inter-element coupling implementation is necessary to prevent scan anomalies due to grating lobes or array resonances. Figure 13 shows data collected from a 32x32 element CSA breadboard designed to operate up to 15 GHz with an element spacing of 0.45”. A centrally-located 32 element line array was scanned using pha shifters at each element, and no issues were obrved over the +/- 45º sc
an range. The mid-band scanned CSA patterns shown are reprentative of the scan performance over the frequency band. Additionally, as array theory predicts, the array scan
loss tracks the embedded element pattern.
Figure 12(a). E-plane and H-plane array patterns at 2 GHz
from breadboard in Figure 9.
Figure 12(b). E-plane and H-plane array patterns at 18
GHz from breadboard in Figure 9.
Figure 13. E-plane and H-plane embedded element and scanned array patterns at 7 GHz from 32x32 CSA
breadboard. A VHF/UHF band CSA breadboard was designed to be
flush-mounted into an 18” x 36” x 10” rectangular cavity.
